Gain variable voltage/current conversion circuit and filter circuit using the same

ABSTRACT

The present invention provides a voltage-current converting circuit which is capable of varying a gain in a wide range without a switching circuit by applying a control voltage to a single control terminal. The voltage-current converting circuit is comprised of a parallel circuit including positive resistors R 1  and R 2,  and transistors Q 3  and Q 4  each acting as a negative resistor, and electrically connected in parallel with the positive resistors R 1  and R 2,  and transistors Q 1  and Q 2  each carrying out voltage-current conversion and electrically connected in series to the parallel circuit. A variable voltage source VV is electrically connected between the transistors Q 3,  Q 4  and a ground. By controlling a voltage of the variable voltage source VV, resistances of the transistors Q 3  and Q 4  are controlled. By varying a voltage of the variable voltage source VV, resistances of the transistors Q 3  and Q 4  vary, resulting in that a voltage between a gate and a source in the transistors Q 1  and Q 2  varies, and thus, a mutual conductance gm of the voltage-current converting circuit varies.

FIELD OF THE INVENTION

The invention relates to a voltage-current conversion circuit (gmamplifier) having a variable conversion gain, and further to a filteringcircuit including a combination circuit comprised of the voltage-currentconversion circuit and a capacity device.

PRIOR ART

In recent years, there is a need for a receiver (multi-mode receiver)designed to operate in a plurality of wireless communication systems.

Such a receiver is required to include a filtering circuit (multi-modefilter) for selecting a channel in accordance with each of wirelesscommunication systems. The filtering circuit is required to have afunction of varying a pass band width in a wide range.

In general, if a receiver is comprised of one chip, there would beselected gm-C system in which a channel-selecting filtering circuit iscomprised of a voltage-current conversion circuit (a gm amplifier) and acapacity device. In order for the channel-selecting filtering circuit tohave a function of varying a pass band width, the voltage-currentconversion circuit is necessary to have a function of varying aconversion gain in a wide range.

A voltage-current conversion circuit (a gm amplifier) is comprisedgenerally of a bipolar transistor, a MOSFET transistor, or other activedevices. A voltage-current conversion circuit is actually designed tohave a mutual conductance (gm value) which is electrically controllablerelative to a designed conductance within ±30%, in order to absorbvariance in a process. In order to control a mutual conductance beyond±30%, there is generally used a switching circuit.

As an example of a voltage-current conversion circuit including aswitching circuit, a MOS type gm amplifier having linearity enhanced bysource degeneration process and further having a widely variable gain issuggested in IEEE, JSSC. Vol. 35, No. 4, pp. 476-489 (April 2000). FIG.23 is a circuit diagram of the suggested MOS type gm amplifier.

The MOS type gm amplifier illustrated in FIG. 23 is comprised of n-typeMOSFET transistors Q21 and Q22 each carrying out voltage-currentconversion, positive resistors R21, R23 and R25 all electricallyconnected in series between a source of the n-type MOSFET transistor Q21and a grounded voltage, positive resistors R22, R24 and R26 allelectrically connected in series between a source of the n-type MOSFETtransistor Q22 and a grounded voltage, a switching circuit SW1electrically connected between a connection node at which the positiveresistors R21 and R23 are electrically connected to each other and aconnection node at which the positive resistors R22 and R24 areelectrically connected to each other, and a switching circuit SW2electrically connected between a connection node at which the positiveresistors R23 and R25 are electrically connected to each other and aconnection node at which the positive resistors R24 and R26 areelectrically connected to each other.

In an operation of the MOS type gm amplifier illustrated in FIG. 23,when an input voltage signal Vin+ is input into a gate of the n-typeMOSFET transistor Q21, there is obtained an output current Iout+, andwhen an input voltage signal Vin− is input into a gate of the n-typeMOSFET transistor Q22, there is obtained an output current Iout−.

FIG. 24 is a circuit diagram of a source degeneration type gm amplifier.

The source degeneration type gm amplifier illustrated in FIG. 24 iscomprised of a n-type MOSFET transistor Q21 carrying out voltage-currentconversion, and a positive resistor R21 electrically connected at oneend to a source of the n-type MOSFET transistor Q21, and at the otherend grounded.

In an operation of the source degeneration type gm amplifier illustratedin FIG. 24, when an input voltage signal Vin is input into a gate of then-type MOSFET transistor Q21, there is obtained an output current Iout.

The MOS type gm amplifier illustrated in FIG. 23 is equivalent to adifferential version of the source degeneration type gm amplifierillustrated in FIG. 24.

Specifically, the MOS type gm amplifier illustrated in FIG. 23 isequivalent to an amplifier in which the n-type MOSFET transistor Q21 ofthe source degeneration type gm amplifier illustrated in FIG. 24 isreplaced with a pair of the n-type MOSFET transistors Q21 and Q22, thepositive resistor R21 is replaced with the positive resistors R21, R23and R25 and the positive resistors R22, R24 and R26, and thecorresponding differential pairs are electrically connected to eachother through the switching circuits SW1 and SW2.

A mutual conductance Gm (Gm=Iout/Vin) in the source degeneration type gmamplifier illustrated in FIG. 24 is expressed with the followingequation (1) wherein gm₀ indicates a mutual conductance of the n-typeMOSFET transistor Q21, and R indicates a resistance of the positiveresistor R21.Gm=gm ₀/(1+gm ₀ ·R)   (1)

The equation (1) indicates that a mutual conductance Gm can becontrolled by varying a resistance of the positive resistor R21.

In the MOS type gm amplifier illustrated in FIG. 23, when the switchingcircuits SW1 and SW2 are off, a resistance between sources of the n-typeMOSFET transistors Q21 and Q22 and a ground is equal to a sum ofresistances of the positive resistors R21, R23 and R25 or a sum ofresistances of the positive resistors R22, R24 and R26.

In contrast, when the switching circuit SW1 is on, since the MOS type gmamplifier illustrated in FIG. 23 is a differential circuit, it can besaid that a node including the switching circuit SW1 is grounded in anAC manner. Accordingly, it can be said that only the positive resistorR21 or R22 is electrically connected in an AC manner between a source ofthe n-type MOSFET transistor Q21 or Q22 and a ground.

That is, the resistance R in the equation (1) is equal to a sum of theresistances of the resistors R21, R23 and R25 (or a sum of theresistances of the resistors R22, R24 and R26) when the switchingcircuits SW1 and SW2 are off, and is equal to the resistance of theresistor R21 (or the resistance of the resistor R22) when the switchingcircuit SW1 is on.

If the resistances of the resistors R21 to R26 are equal to one another,and the mutual conductance gm₀ of the n-type MOSFET transistor Q21 isequal to 1/(the resistance of the resistor R21), the MOS type gmamplifier illustrated in FIG. 23 would have a double-variable mutualconductance Gm.

The MOS type gm amplifier illustrated in FIG. 23 is characterized inthat since bias voltages remain unchanged at nodes, even if theswitching circuits SW1 and SW2 are turned on or off to change currentpaths, the mutual conductance gm₀ in the equation (1) can be treated asa fixed conductance, and accordingly, a mutual conductance Gm can bevaried only by controlling the resistances of the resistors.

FIG. 25 is a circuit diagram of a MOS type gm amplifier as second priorart, disclosed in IEEE. JSSC. Vol. 37, No. 2, pp. 125-136, February2002. FIG. 25(a) is a circuit diagram of the entirety, and FIG. 25(b) isa circuit diagram of the programmable current mirror circuits G1 and G2found in FIG. 25(a).

The MOS type gm amplifier illustrated in FIG. 25(a) is comprised ofp-type MOSFET transistors Q23, Q24, Q25 and Q26, current sources CS1,CS2 and CS3, a voltage source VS, and programmable current mirrorcircuits G1 and G2.

The current source CS1 is electrically connected to the voltage sourceVS and drains of the p-type MOSFET transistors Q23 and Q26. The voltagesource VS is electrically connected to drains of the p-type MOSFETtransistors Q24 and Q25. The p-type MOSFET transistors Q23 and Q25 areelectrically connected through sources thereof to the programmablecurrent mirror circuit G1, and the p-type MOSFET transistors Q24 and Q26are electrically connected through sources thereof to the programmablecurrent mirror circuit G2. The current source CS2 is electricallyconnected to the programmable current mirror circuit G1, and the currentsource CS3 is electrically connected to the programmable current mirrorcircuit G2. The p-type MOSFET transistors Q23 and Q24 receive an inputvoltage signal Vin+ through gates thereof, and the p-type MOSFETtransistors Q25 and Q26 receive an input voltage signal Vin− throughgates thereof.

Each of the programmable current mirror circuits G1 and G2 illustratedin FIG. 25(b) is comprised of n-type MOSFET transistors Q27, Q28, Q29,Q30, Q31, Q32, Q33, Q34, Q35 and Q36, and switching circuits SW3, SW4and SW5 electrically connected to gates of the n-type MOSFET transistorsQ31, Q32, and Q33.

The programmable current mirror circuits G1 and G2 are designed to havesuch a structure that the n-type MOSFET transistors Q31, Q32 and Q33through which a current output from the MOS type gm amplifier runs arearranged in parallel, and a MOSFET transistor which actually operatescan be selected among the n-type MOSFET transistors Q31, Q32 and Q33 bymeans of the switching circuits SW3, SW4 and SW5.

On entry of differential input voltage signals Vin+ and Vin− into gatesof the p-MOSFET transistors Q23, Q24 and Q25, Q26, respectively, acurrent having differential components associated with the differentialinput voltage runs into the programmable current mirror circuits G1 andG2 through the MOSFET transistors Q23, Q24, Q25 and Q26. By turning onor off the switching circuits SW3 to SW5, the differential componentsare amplified by desired times, and a desired current is output from theprogrammable current mirror circuits G1 and G2.

In a condition illustrated in FIG. 25, since the switching circuits SW3and SW4 in the programmable current mirror circuits G1 and G2 have apath to the voltage source, the n-type MOSFET transistors Q31 and Q32are on. In order to reduce the mutual conductance Gm, a path of theswitching circuit SW4 is turned to a ground. As a result, the n-typeMOSFET transistor Q32 is turned off, and thus, the mutual conductance Gmis reduced. In order to increase the mutual conductance Gm, a path ofthe switching circuit SW5 is turned to the voltage source. As a result,the n-type MOSFET transistor Q33 is turned on, and thus, the mutualconductance Gm is increased.

The programmable current mirror circuits G1 and G2 illustrated in FIG.25(b) are characterized in that since the switching circuits SW3, SW4and SW5 are electrically connected at one ends thereof to gates of then-type MOSFET transistors Q31, Q32 and Q33, respectively, the circuitsG1 and G2 are less influenced by parasitic factors (resistance,capacity, and so on) of the switching circuits SW3, SW4 and SW5.

The greater the number of MOSFET transistors arranged in parallel is,the greater a variable range of the mutual conductance Gm is.

The above-mentioned prior art was necessary to have a switchingcircuit(s) in order to widen a variable range of a gain of avoltage-current converting circuit (gm amplifier). Hence, the prior artwas necessary to include a digital circuit, resulting in that a circuitstructure was unavoidably complex, and an area of a chip was unavoidablyincreased.

The first prior art circuit illustrated in FIG. 23 is accompanied with aproblem that since a current runs through the switching circuits SW1 andSW2, the circuit is much influenced by a parasitic impedance of theswitching circuits SW1 and SW2.

The second prior art circuit illustrated in FIG. 25 is accompanied witha problem that a lot of MOSFET transistors acting as a current sourcehave to be arranged in parallel for ensuring a wide variable range of again, and if only a minimum number of MOSFET transistors operate, acapacity of the rest of MOSFET transistors, that is, non-operatingMOSFET transistors causes harmful influences. Hence, when a filterhaving a variable pass band is comprised of the voltage-currentconverting circuit (gm amplifier) of the second prior art, the filterunavoidably has to have a complex structure, and a chip is unavoidablylarge in size.

Japanese Patent Application Publication No. 3-64109 has suggested adifferentially amplifying circuit including a pair of MOSFET transistorsfor increasing a mutual conductance of differentially amplifying stages.Source electrodes of the pair of MOSFET transistors are electricallyconnected to each other through nodes, and active devices areelectrically connected between the source electrodes and the nodes.Thus, a function of a negative resistance device is accomplished.

Japanese Patent Application Publication No. 7-235840 has suggested anamplifying circuit having a variable gain, comprised of a first pair oftransistors receiving an input through bases thereof, a PN junction pairreceiving a collector current from each of the first pair oftransistors, as a bias current, a second pair of transistors including acurrent provider which provides a current to a common emitter receivinga voltage difference in the PN junction pair as an input through a basethereof, and a third pair of transistors including a current providerhaving collector current paths electrically connected to emitters of thefirst pair of transistors, and bases electrically connected to thecollector current paths, the emitters being electrically connected toeach other through an impedance, the current provider providing a biascurrent to the emitters. A collector of the second pair of transistorsprovides an output.

Japanese Patent Application Publication No. 2001-36356 has suggested avoltage-current converting circuit comprised of a first circuitincluding a MOS transistor differential pair, a second circuit includinga MOS transistor differential pair having drain terminals electricallyconnected to source terminals of the first circuit, and a resistorelectrically connected between sources of the second circuit. Gateterminals of the first circuit act as an input voltage terminal, anddrain terminals of the first circuit act as an output current terminal.Each of gates of two MOS transistors complementary to each other in thesecond circuit is electrically connected to a drain of the other MOStransistor, and sources of the two MOS transistors are grounded througha current source.

However, the above-mentioned problems remain unsolved even in thecircuits suggested in the above-mentioned Publications.

In view of the above-mentioned problems in the prior art, a first objectof the present invention is to provide a voltage-current convertingcircuit which is capable of varying a gain in a wide range without aswitching circuit, a second object of the present invention is tosimplify a circuit structure to thereby reduce an area of a chip, and athird object of the present invention is to accomplish a filter having ahighly variable range of a pass band, with the simplified circuitstructure for accomplishing a multi-mode receiver having a small area ofa chip.

DISCLOSURE OF THE INVENTION

In order to accomplish the above-mentioned objects, the presentinvention provides a voltage-current converting circuit which outputs acurrent in accordance with a voltage input thereto, including an activedevice having an input terminal, an output terminal, and a groundedterminal, and carrying out voltage-current conversion, and a resistorcircuit electrically connected in series to the active device throughthe grounded terminal of the active device, and controlling a conversiongain of the active device, the resistor circuit having a variableresistance, and including a negative resistance device.

In the voltage-current converting circuit in accordance with the presentinvention, the active device carrying out voltage-current conversion iselectrically connected in series to the resistor circuit including anegative resistance device and having a variable resistance. Forinstance, by designing the negative resistance device to have a variableresistance, it would be possible to much vary a resistance of theresistor circuit. Hence, it would be possible for the active device tohave a broad variable range of a voltage-current conversion gain. Forinstance, the negative resistance device may be comprised of a MOSFETtransistor or a bipolar transistor. Accordingly, since it is possible tocontrol the resistance by a single control signal, that is, by applyinga control voltage to a single control terminal, the voltage-currentconversion circuit can be compactly formed of a small number of circuitelements without necessity of using a switching circuit. In addition, acombination of the voltage-current converting circuit and a capacitydevice could present a filter having a simple circuit structure, buthaving a broad variable range of a pass band.

In the voltage-current converting circuit in accordance with the presentinvention, the active device may be comprised of a pair of activedevices each operating differentially with each other, and each havingan input terminal, an output terminal, and a grounded terminal, andcarrying out voltage-current conversion, and the resistor circuit may becomprised of a pair of resistor circuits each electrically connected inseries to each of the active devices through the grounded terminal ofeach of the active devices, and each controlling a conversion gain ofeach of the active devices, each of the resistor circuits having avariable resistance, and including a negative resistance device.

It is preferable that the negative resistance device has a variableresistance.

The resistor circuit may be designed to have various structures, asmentioned below.

For instance, the resistor circuit or each of the resistor circuits maybe comprised of one or a plurality of resistance device(s) electricallyconnected in series to the active device, and a negative resistancedevice electrically connected in parallel with at least one of theresistance device(s).

The resistor circuit or each of the resistor circuits may be comprisedof a first circuit comprised of a resistance device and a negativeresistance device electrically connected in series to each other, thefirst circuit being electrically connected in series to the activedevice.

The resistor circuit or each of the resistor circuits may be comprisedof a first resistance device electrically connected in series to theactive device, and a second circuit electrically connected in parallelwith the first resistance device, the second circuit being comprised ofa negative resistance device, and a second resistance deviceelectrically connected in series to the negative resistance device.

It is preferable that the negative resistance device of the pair ofresistance circuits is comprised of a pair of active deviceselectrically connected in cross to each other and operatingdifferentially with each other, and each receiving, as an input signal,a node signal either at a connection node at which the active device andthe resistor circuit are electrically connected to each other or at anyconnection node in the resistor circuit.

For instance, the negative resistance device is comprised of a fieldeffect transistor or a bipolar transistor.

A resistance of the negative resistance device may be controlled bycontrolling either a source voltage or an emitter voltage of the fieldeffect transistor or bipolar transistor.

It is preferable that the voltage-current converting circuit inaccordance with the present invention further includes avoltage-providing circuit electrically connected between a referencevoltage point and either a source or an emitter of the field effecttransistor or bipolar transistor, and wherein a resistance of thenegative resistance device is controlled by controlling a voltageprovided by the voltage-providing circuit.

For instance, the voltage-providing circuit may be comprised of anoperational amplifier having a first input terminal, a second inputterminal, and an output terminal, and an active device. Avoltage-control signal is input to the first input signal of theoperational amplifier, an input terminal of the active device iselectrically connected to the output terminal of the operationalamplifier, and an output terminal of the active device is electricallyconnected to the second input terminal of the operational amplifier.

It is preferable that the negative resistance device is comprised of apair of field effect transistors or bipolar transistors operatingdifferentially with each other, wherein sources or emitters of the fieldeffect transistors or bipolar transistors are electrically connected toeach other.

The voltage-current converting circuit in accordance with the presentinvention may further include a voltage-controller electricallyconnected to a connection node at which the active device and theresistor circuit are electrically connected to each other, forcontrolling a voltage of the connection node.

The voltage-controller may be comprised of an active device electricallyconnected between a reference voltage and the connection node, andhaving an input terminal to which a bias signal is input.

For instance, the voltage-controller may be designed to compensate forvoltage fluctuation caused at the connection node by variance of aresistance of the negative resistance device.

The resistor circuit may be designed to include a variable resistorhaving a positive resistance.

The variable resistor may be comprised of an active device.

The active device may be comprised of a field effect transistor or abipolar transistor.

The active device carrying out voltage-current conversion and the activedevice comprising the negative resistance device may be comprised of thesame transistors having electrical conductivities different from eachother.

The present invention further provides a filtering circuit including acombination circuit comprised of a voltage-current converting circuit,and a capacity device. As the voltage-current converting circuit is usedthe above-mentioned voltage-current converting circuit, ensuring that apass band can be controlled by varying a gain of the voltage-currentconverting circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1(a) is a circuit diagram of a voltage-current converting circuitin accordance with the first embodiment of the present invention, andFIG. 1(b) is used for explanation of an operation of the voltage-currentconverting circuit.

FIG. 2(a) is a circuit diagram of a variant of a voltage-currentconverting circuit in accordance with the first embodiment of thepresent invention, and FIG. 2(b) is used for explanation of an operationof the variant.

FIG. 3(a) is a circuit diagram of a voltage-current converting circuitin accordance with the second embodiment of the present invention, andFIG. 3(b) is used for explanation of an operation of the voltage-currentconverting circuit.

FIG. 4(a) is a circuit diagram of a voltage-current converting circuitin accordance with the third embodiment of the present invention, andFIG. 4(b) is used for explanation of an operation of the voltage-currentconverting circuit.

FIG. 5(a) is a circuit diagram of a voltage-current converting circuitin accordance with the fourth embodiment of the present invention, andFIG. 5(b) is used for explanation of an operation of the voltage-currentconverting circuit.

FIG. 6(a) is a circuit diagram of a voltage-current converting circuitin accordance with the fifth embodiment of the present invention, andFIG. 6(b) is used for explanation of an operation of the voltage-currentconverting circuit.

FIG. 7 is a circuit diagram of an example of the variable voltage sourcein the fifth embodiment of the present invention.

FIG. 8(a) is a circuit diagram of a voltage-current converting circuitin accordance with the sixth embodiment of the present invention, andFIG. 8(b) is used for explanation of an operation of the voltage-currentconverting circuit.

FIG. 9 is a circuit diagram of an example of the bias circuit in thesixth embodiment of the present invention.

FIG. 10(a) is a circuit diagram of a voltage-current converting circuitin accordance with the seventh embodiment of the present invention, andFIG. 10(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 11(a) is a circuit diagram of a voltage-current converting circuitin accordance with the eighth embodiment of the present invention, andFIG. 11(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 12(a) is a circuit diagram of a voltage-current converting circuitin accordance with the ninth embodiment of the present invention, andFIG. 12(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 13 is a circuit diagram of an example of the phase-invertingcircuit in the ninth embodiment of the present invention.

FIG. 14(a) is a circuit diagram of a voltage-current converting circuitin accordance with the tenth embodiment of the present invention, andFIG. 14(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 15 is a circuit diagram of a first example of the variable positiveresistor in the tenth embodiment of the present invention.

FIG. 16 is a circuit diagram of a second example of the variablepositive resistor in the tenth embodiment of the present invention.

FIG. 17(a) is a circuit diagram of a voltage-current converting circuitin accordance with the eleventh embodiment of the present invention, andFIG. 17(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 18(a) is a circuit diagram of a voltage-current converting circuitin accordance with the twelfth embodiment of the present invention, andFIG. 18(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 19(a) is a circuit diagram of a voltage-current converting circuitin accordance with the thirteenth embodiment of the present invention,and FIG. 19(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 20(a) is a circuit diagram of a voltage-current converting circuitin accordance with the fourteenth embodiment of the present invention,and FIG. 20(b) is used for explanation of an operation of thevoltage-current converting circuit.

FIG. 21(a) is a circuit diagram of a filter circuit in accordance withthe fifteenth embodiment of the present invention, and FIG. 21(b) is acircuit diagram of the voltage-current converting circuit used in thefilter circuit.

FIG. 22 is used for explanation of an operation of the filter circuit inaccordance with the fifteenth embodiment of the present invention.

FIG. 23 is a circuit diagram of the MOS type gm amplifier as the firstprior art.

FIG. 24 is a circuit diagram of the source degeneration type gmamplifier.

FIG. 25(a) is a circuit diagram of the MOS type gm amplifier as thesecond prior art, and FIG. 25(b) is a circuit diagram of a programmablecurrent mirror circuit used in the MOS type gm amplifier.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS First Embodiment

FIG. 1(a) is a circuit diagram of a voltage-current converting circuitin accordance with the first embodiment of the present invention, andFIG. 1(b) is used for explanation of an operation of the voltage-currentconverting circuit.

The voltage-current converting circuit in accordance with the firstembodiment is comprised of an n-type MOSFET transistor Q0 acting as anactive device carrying out voltage-current conversion, and a resistorcircuit electrically connected in series to the n-type MOSFET transistorQ0. The resistor circuit is comprised of a positive resistor R0electrically connected in series to the n-type MOSFET transistor Q0 andgrounded, and a negative resistor NR electrically connected in series tothe n-type MOSFET transistor Q0, electrically connected in parallel withthe positive resistor R0, grounded, and having a variable resistance.

On entry of an input voltage signal Vin into a gate of the n-type MOSFETtransistor Q0, there is obtained an output current Iout.

Hereinbelow is explained a principle of an operation of thevoltage-current converting circuit (gm amplifier) in accordance with thefirst embodiment.

A mutual conductance Gm (Gm=Iout/Vout) of the voltage-current convertingcircuit in accordance with the first embodiment is expressed with theequation (2) which is obtained by replacing R with 1/(1/R₀−1/R_(NR)) inthe equation (1). $\begin{matrix}{{Gm} = {\frac{1}{1 + {\frac{1}{\frac{1}{R_{0}} - \frac{1}{R_{NR}}}{gm}_{0}}}{gm}_{0}}} & (2)\end{matrix}$

In the equation (2), R0 indicates a resistance of the positive resistorR₀, R_(NR) indicates an absolute value of a resistance of the negativeresistor NR, and gm₀ indicates a mutual conductance gm of the n-typeMOSFET transistor Q0.

FIG. 1(b) is a graph showing how the mutual conductance Gm of thevoltage-current converting circuit varies when the resistance R_(NR)varies in the equation (2).

As shown with a solid line 101 in FIG. 1(b), when the resistance R_(NR)of the negative resistor NR varies from R0 to infinity, the mutualconductance Gm can be varied from 0 to (gm₀/(1+gm₀·R)). That is, it ispossible to vary the mutual conductance Gm to infinity.

As shown with a solid line 102 in FIG. 1(b), when the resistance R_(NR)varies from (R₀/(1+gm₀·R₀)) to R₀, the mutual conductance Gm can bevaried from minus infinity to zero. That is, it is possible to vary themutual conductance Gm to minus infinity.

As shown with a solid line 103 in FIG. 1(b), when the resistance R_(NR)varies from zero to (R₀/(1+gm₀·R₀)), the mutual conductance Gm can bevaried from (gm₀/(1+gm₀·R)) to infinity. That is, it is possible to varythe mutual conductance Gm to infinity. If R₀ is set equal to 1/gm₀(R₀=1/gm₀), it is possible to vary the mutual conductance Gm from gm₀/2to infinity.

It should be noted that when the resistance R_(NR) is varied from(R₀/(1+gm₀·R₀)) to R₀, the mutual conductance Gm would be negative, andthe output current Iout would run in an opposite direction in comparisonwith other cases. The first embodiment contains cases in which themutual conductance Gm is negative.

In the voltage-current converting circuit in accordance with the firstembodiment, it is not always necessary to vary the resistance R_(NR) ofthe negative resistor NR in a wide range. A range in which theresistance R_(NR) varies can be determined in accordance with anecessary variable range of the mutual conductance Gm. For instance, itis possible to vary the resistance R_(NR) of the negative resistor NRonly in a finite range within a range from R₀ to infinity.

In the voltage-current converting circuit in accordance with the firstembodiment, illustrated in FIG. 1, a number of a resistor device (thepositive resistor R0) electrically connected in series to the n-typeMOSFET transistor Q0 acting as an active device is one, but two or moreresistor devices may be electrically connected in series to the n-typeMOSFET transistor Q0. An example thereof is illustrated in FIG. 2 as avariant of the first embodiment.

FIG. 2(a) is a circuit diagram of a variant of the voltage-currentconverting circuit in accordance with the first embodiment of thepresent invention, and FIG. 2(b) is used for explanation of an operationof the variant.

The voltage-current converting circuit in accordance with the variant ofthe first embodiment is comprised of an n-type MOSFET transistor Q0acting as an active device carrying out voltage-current conversion, anda resistor circuit electrically connected in series to the n-type MOSFETtransistor Q0. The resistor circuit is comprised of a positive resistorR00 electrically connected in series to the n-type MOSFET transistor Q0,a positive resistor R0 electrically connected in series to the positiveresistor R00 and grounded, and a negative resistor NR electricallyconnected in series to the positive resistor R00, electrically connectedin parallel with the positive resistor R0, grounded, and having avariable resistance.

Similarly to the voltage-current converting circuit in accordance withthe first embodiment, on entry of an input voltage signal Vin into agate of the n-type MOSFET transistor Q0, there is obtained an outputcurrent Iout.

A mutual conductance Gm of the voltage-current converting circuit inaccordance with the variant of the first embodiment is expressed withthe equation (5) which is obtained by replacing R with(R₀₀+1/(1/R₀−1/R_(NR))) in the equation (1). $\begin{matrix}{{Gm} = {\frac{1}{1 + {\left( {R_{00} + \frac{1}{\frac{1}{R_{0}} - \frac{1}{R_{NR}}}} \right){gm}_{0}}}{gm}_{0}}} & (5)\end{matrix}$

FIG. 2(b) is a graph showing how the mutual conductance Gm of thevoltage-current converting circuit varies when the resistance R_(NR) ofthe negative resistor NR varies in the equation (5).

As shown with a solid line 201 in FIG. 2(b), when the resistance R_(NR)is equal to the resistance R₀(R_(NR)=R₀), the mutual conductance Gm isequal to zero. When the resistance R_(NR) is equal to infinity, themutual conductance Gm is equal to (gm₀/(1+(R₀₀+R₀)·gm₀)). That is, themutual conductance Gm can have an infinite variable range. By settingthe resistances R₀ and R₀₀ equal to 1/gm₀(R₀=R₀₀=1/gm₀), the mutualconductance Gm is equal to gm₀/3.

As shown with a solid line 202 in FIG. 2(b), when the resistance R_(NR)varies from (R₀(1+gm₀·R₀₀)/(1+(R₀₀+R₀)gm₀)) to R₀, the mutualconductance Gm can be varied from minus infinity to zero. That is, it ispossible to vary the mutual conductance Gm to minus infinity.

As shown with a solid line 203 in FIG. 2(b), when the resistance R_(NR)varies from zero to (R₀(1+gm₀·R₀₀)/(1+(R₀₀+R₀)gm₀)), the mutualconductance Gm can be varied from (gm₀/(1+(R₀₀+R₀) gm₀)) to infinity.That is, it is possible to vary the mutual conductance Gm to infinity.

As mentioned above, a plurality of resistors may be electricallyconnected in series to the n-type MOSFET transistor Q0, in which case,the negative resistance device NR is electrically connected in parallelwith at least one of the resistors.

Second Embodiment

FIG. 3(a) is a circuit diagram of a voltage-current converting circuitin accordance with the second embodiment of the present invention, andFIG. 3(b) is used for explanation of an operation of the voltage-currentconverting circuit.

The voltage-current converting circuit in accordance with the secondembodiment is comprised of an n-type MOSFET transistor Q0 acting as anactive device carrying out voltage-current conversion, and a resistorcircuit electrically connected in series to the n-type MOSFET transistorQ0. The resistor circuit is comprised of a negative resistor NRelectrically connected in series to the n-type MOSFET transistor Q0, andhaving a variable resistance, and a positive resistor R0 electricallyconnected in series to the negative resistor NR and grounded.

A mutual conductance Gm of the voltage-current converting circuit inaccordance with the second embodiment is expressed with the equation (3)which is obtained by replacing R with (R₀−R_(NR)) in the equation (1).$\begin{matrix}{{Gm} = {\frac{1}{1 + {\left( {R_{0} - R_{NR}} \right){gm}_{0}}}{gm}_{0}}} & (3)\end{matrix}$

FIG. 3(b) is a graph showing how the mutual conductance Gm of thevoltage-current converting circuit varies when the resistance R_(NR) ofthe negative resistor NR varies in the equation (3).

As shown with a solid line 301 in FIG. 3(b), when the resistance R_(NR)is equal to infinity, the mutual conductance Gm is equal to zero (Gm=0).When the resistance R_(NR) is equal to (R₀+1/gm₀), the mutualconductance Gm is equal to minus infinity (Gm=−∞). That is, the mutualconductance Gm can have an infinite variable range.

As shown with a solid line 302 in FIG. 3(b), when the resistance R_(NR)varies from zero to (R₀+1/gm₀), the mutual conductance Gm can be variedfrom zero to infinity. That is, it is possible to vary the mutualconductance Gm to infinity.

Third Embodiment

FIG. 4(a) is a circuit diagram of a voltage-current converting circuitin accordance with the third embodiment of the present invention, andFIG. 4(b) is used for explanation of an operation of the voltage-currentconverting circuit.

The voltage-current converting circuit in accordance with the thirdembodiment is comprised of an n-type MOSFET transistor Q0 acting as anactive device carrying out voltage-current conversion, and a resistorcircuit electrically connected in series to the n-type MOSFET transistorQ0. The resistor circuit is comprised of a positive resistor R0, as afirst resistance device, electrically connected in series to the n-typeMOSFET transistor Q0 and grounded, and a second resistor circuitelectrically connected in parallel with the positive resistor R0. Thesecond resistor circuit is comprised of a negative resistor NRelectrically connected in series to the n-type MOSFET transistor Q0, andhaving a variable resistance, and a positive resistor R00, as a secondresistance device, electrically connected in series to the negativeresistor NR, and grounded.

A mutual conductance Gm of the voltage-current converting circuit inaccordance with the third embodiment is expressed with the equation (4)which is obtained by replacing R with 1/(1/R₀−1/(R_(NR)−R₀₀)) in theequation (1). R₀₀ indicates a resistance of the positive resistor R00.$\begin{matrix}{{Gm} = {\frac{1}{1 + {\left( \frac{1}{\frac{1}{R_{0}} - \frac{1}{R_{NR} - R_{00}}} \right){gm}_{0}}}{gm}_{0}}} & (4)\end{matrix}$

FIG. 4(b) is a graph showing how the mutual conductance Gm of thevoltage-current converting circuit varies when the resistance R_(NR) ofthe negative resistor NR varies in the equation (4).

As shown with a solid line 401 in FIG. 4(b), when the resistance R_(NR)is equal to R₀+R₀₀, the mutual conductance Gm is equal to zero (Gm=0).When the resistance R_(NR) is equal to infinity, the mutual conductanceGm is equal to (gm₀/(1+gm₀·R)). That is, the mutual conductance Gm canhave an infinite variable range. By setting the resistances R₀ of thepositive resistor R0 equal to 1/gm₀ (R₀=1/gm₀), the mutual conductanceGm is equal to gm₀/2.

As shown with a solid line 402 in FIG. 4(b), when the resistance R_(NR)varies from (R₀₀+R₀/(1+R₀ gm₀) to R₀+R₀₀, the mutual conductance Gm canbe varied from minus infinity to zero. That is, it is possible to varythe mutual conductance Gm to minus infinity.

As shown with a solid line 403 in FIG. 4(b), when the resistance R_(NR)varies from zero to (R₀₀+R₀/(1+R₀ gm₀), the mutual conductance Gm can bevaried from (gm₀/(1+gm₀·R)) to infinity. That is, it is possible to varythe mutual conductance Gm to infinity.

Fourth Embodiment

FIG. 5(a) is a circuit diagram of a voltage-current converting circuitin accordance with the fourth embodiment of the present invention, andFIG. 5(b) is used for explanation of an operation of the voltage-currentconverting circuit.

The voltage-current converting circuit in accordance with the fourthembodiment is comprised of an n-type MOSFET transistor Q0 acting as anactive device carrying out voltage-current conversion, and a resistorcircuit electrically connected in series to the n-type MOSFET transistorQ0. The resistor circuit is comprised only of a negative resistor NRhaving a variable resistance.

A mutual conductance Gm of the voltage-current converting circuit inaccordance with the fourth embodiment is expressed with the equation (6)which is obtained by replacing R with (−R_(NR)) in the equation (1).$\begin{matrix}{{Gm} = {\frac{1}{1 - {R_{NR} \cdot {gm}_{0}}}{gm}_{0}}} & (6)\end{matrix}$

FIG. 5(b) is a graph showing how the mutual conductance Gm of thevoltage-current converting circuit varies when the resistance R_(NR) ofthe negative resistor NR varies in the equation (6).

As shown with a solid line 501 in FIG. 5(b), when the resistance R_(NR)is equal to (1/gm₀), the mutual conductance Gm is equal to minusinfinity. When the resistance R_(NR) is equal to infinity, the mutualconductance Gm is equal to zero (Gm=0). That is, the mutual conductanceGm can have an infinite variable range.

As shown with a solid line 502 in FIG. 5(b), when the resistance R_(NR)varies from zero to (1/gm₀), the mutual conductance Gm can be variedfrom zero to infinity. That is, it is possible to vary the mutualconductance Gm to infinity.

Though a n-type MOSFET transistor is used as an active device forcarrying out voltage-current conversion in the above-mentioned first tofourth embodiments, it should be noted that any active device such as abipolar transistor or a MESFET may be used in place of a n-type MOSFETtransistor.

Though the negative resistor NR is designed to have a variableresistance in the above-mentioned first to fourth embodiments, it shouldbe noted that the negative resistor may be designed to have a fixedresistance, and the positive resistors R0 and R00 may be designed tohave a variable resistance.

For instance, in the voltage-current converting circuit in accordancewith the first embodiment, illustrated in FIG. 1(a), if the positiveresistor R0 is designed to have a variable resistance, it would bepossible to vary the mutual conductance Gm from zero to infinity byvarying the resistance R₀ from R_(NR) to infinity in the equation (2),assuming that R_(NR) is equal to 1/gm₀. Thus, it is possible to vary themutual conductance Gm to infinity. Those negative or positive variableresistors can be comprised of an active device such as a MOSFETtransistor.

In the above-mentioned first to fourth embodiments, two active deviceseach carrying out voltage-current conversion may be connected in crossto each other such that they differentially operate, and complementaryinput voltages may be input into the two active devices to havecomplementary output currents. Hereinbelow are explained embodiments inwhich two active devices are electrically connected in cross to eachother such that they differentially operate.

Fifth Embodiment

FIG. 6(a) is a circuit diagram of a voltage-current converting circuitin accordance with the fifth embodiment of the present invention.

The voltage-current converting circuit in accordance with the fifthembodiment is comprised of n-type MOSFET transistors Q1 and Q2 each asan active device carrying out voltage-current conversion, positiveresistors R1 and R2 electrically connected in series to the n-typeMOSFET transistors Q1 and Q2, respectively, and grounded, a resistorcircuit electrically connected between a junction node at which then-type MOSFET transistor Q1 and the positive resistor R1 areelectrically connected to each other and a junction node at which then-type MOSFET transistor Q2 and the positive resistor R2 areelectrically connected to each other, and a variable voltage source VVelectrically connected in series to the resistor circuit and grounded.

The resistor circuit is comprised of n-type MOSFET transistors Q3 and Q4having the same size as each other and acting as a negative resistor.

The n-type MOSFET transistor Q3 has a gate electrically connected to ajunction node at which the n-type MOSFET transistor Q2 and the positiveresistor R2 are electrically connected to each other, a drainelectrically connected to a junction node at which the n-type MOSFETtransistor Q1 and the positive resistor R1 are electrically connected toeach other, and a source electrically connected to the variable voltagesource VV.

The n-type MOSFET transistor Q4 has a gate electrically connected to ajunction node at which the n-type MOSFET transistor Q1 and the positiveresistor R1 are electrically connected to each other, a drainelectrically connected to a junction node at which the n-type MOSFETtransistor Q2 and the positive resistor R2 are electrically connected toeach other, and a source electrically connected to the variable voltagesource VV.

The n-type MOSFET transistors Q1 and Q2 have the same size as eachother, and output currents Iout+ and Iout− on receipt of input voltagesignals Vin+ and Vin− through the gates thereof, respectively. Thepositive resistors R1 and R2 have the same resistances as each other.

In a MOSFET transistor circuit having a grounded source, a source, adrain and a gate correspond to a grounded terminal, an output terminaland a control terminal, respectively. The positive resistors R1 and R2and the n-type MOSFET transistors Q3 and Q4 are all electricallyconnected to sources or grounded terminals of the n-type MOSFETtransistors Q1 and Q2.

Hereinbelow is explained a principle of an operation of thevoltage-current converting circuit (gm amplifier) in accordance with thefifth embodiment.

Since the voltage-current converting circuit in accordance with thefifth embodiment is equivalent to a circuit having the same structure asthat of the voltage-current converting circuit in accordance with thefirst embodiment, illustrated in FIG. 1, except the negative resistor NRis replaced with the n-type MOSFET transistor Q3, the resistance R_(NR)is equal to 1/gm_(Q3). Accordingly, a mutual conductance Gm(Gm=(Iout+Iout−)/(Vin+−Vin−)) of the voltage-current converting circuitin accordance with the fifth embodiment is expressed with the equation(7) which is obtained by replacing R with 1/(1/R_(R1)−gm_(Q3)) in theequation (1). $\begin{matrix}{{Gm} = {\frac{1}{1 + {\frac{1}{\frac{1}{R_{R1}} - {gm}_{Q3}}{gm}_{0}}}{gm}_{0}}} & (7)\end{matrix}$

In the equation (7), R_(R1) indicates a resistance of the positiveresistors R1 and R2, gm_(Q3) indicates a mutual conductance gm of then-type MOSFET transistors Q3 and Q4, and gm₀ indicates a mutualconductance gm of the n-type MOSFET transistors Q1 and Q2.

As is obvious in view of the equation (7), it is possible to vary themutual conductance Gm from zero to (gm₀/(1+gm₀·R_(R1))) by varying themutual conductance gm_(Q3) of the n-type MOSFET transistors Q3 and Q4from 1/R_(R1) to zero. That is, the mutual conductance Gm can have aninfinite variable range.

The mutual conductance gm_(Q3) of the n-type MOSFET transistors Q3 andQ4 can be controlled in light of the fact that the mutual conductance Gmis in proportion with a voltage Vgs between a gate and a source.Specifically, the voltage Vgs between a gate and a source of the n-typeMOSFET transistors Q3 and Q4 is controlled by varying a voltage of thevariable voltage source VV electrically connected to the sources of then-type MOSFET transistors Q3 and Q4.

For instance, by designing the n-type MOSFET transistors Q3 and Q4 suchthat the mutual conductance gm_(Q3) of the n-type MOSFET transistors Q3and Q4 has a maximum 1/R_(R1) when a voltage of the variable voltagesource VV is in maximum, the mutual conductance gm_(Q3) of the n-typeMOSFET transistors Q3 and Q4 becomes zero when a voltage of the variablevoltage source VV is raised up to a drain voltage of the n-type MOSFETtransistors Q3 and Q4. Hence, the mutual conductance Gm of thevoltage-current converting circuit in accordance with the fifthembodiment varies from zero to (gm₀/(1+gm₀·R_(R1))). That is, the mutualconductance Gm can have an infinite variable range.

FIG. 7 is a circuit diagram of an example of the variable voltage sourceVV.

In FIG. 7, the n-type MOSFET transistors Q3 and Q4 acting as a negativeresistance device in the voltage-current converting circuit inaccordance with the fifth embodiment, illustrated in FIG. 6, are alsoillustrated.

The variable voltage source VV illustrated in FIG. 7 is comprised of anoperational amplifier OA having a first input terminal (minus (−)terminal), a second input terminal (plus (+) terminal), and an outputterminal, and an n-type MOSFET transistor Q5 acting as an active device.A voltage-control signal is input into the first input terminal (minus(−) terminal) of the operational amplifier OA. An input terminal (gate)of the n-type MOSFET transistor Q5 is electrically connected to theoutput terminal of the operational amplifier OA, an output terminal(drain) of the n-type MOSFET transistor Q5 is electrically connected tothe second input terminal (plus (+) terminal) of the operationalamplifier OA, and a ground terminal (source) is grounded.

The n-type MOSFET transistor Q5 acts as a voltage source. Byelectrically connecting a drain voltage of the n-type MOSFET transistorQ5 to the second input terminal (plus (+) terminal) of the operationalamplifier OA, and electrically connecting an output terminal of theoperational amplifier OA to a gate of the n-type MOSFET transistor Q5, acontrol voltage input into the first input terminal (minus (−) terminal)of the operational amplifier OA can be applied to a drain voltage of then-type MOSFET transistor Q5, that is, a source voltage of the n-typeMOSFET transistors Q3 and Q4.

Furthermore, since the n-type MOSFET transistors Q3 and Q4 operatedifferentially each other, a current running through a drain of then-type MOSFET transistor Q5 has no AC components. Accordingly, theoperational amplifier OA is not required to operate in a high-frequencyband, ensuring that the variable voltage source VV illustrated in FIG. 7can act as a stable voltage source.

Sixth Embodiment

FIG. 8 is a circuit diagram of a voltage-current converting circuit inaccordance with the sixth embodiment of the present invention.

The voltage-current converting circuit in accordance with the sixthembodiment additionally includes p-type MOSFET transistors Q6 and Q7,and a bias circuit 1 in comparison with the voltage-current convertingcircuit in accordance with the fifth embodiment, illustrated in FIG. 6.Hence, parts or elements in FIG. 8 that correspond to those of FIG. 6have been provided with the same reference numerals.

The p-type MOSFET transistor Q6 has a source electrically connected to asource of the n-type MOSFET transistor Q1, the positive resistor R1, adrain of the n-type MOSFET transistor Q3, and a gate of the n-typeMOSFET transistor Q4, and a gate electrically connected to the biascircuit 1. The p-type MOSFET transistor Q7 has a source electricallyconnected to a source of the n-type MOSFET transistor Q2, the positiveresistor R2, a drain of the n-type MOSFET transistor Q4, and a gate ofthe n-type MOSFET transistor Q3, and a gate electrically connected tothe bias circuit 1. The bias circuit 1 applies a bias voltage to gatesof the p-type MOSFET transistors Q6 and Q7.

In the voltage-current converting circuit in accordance with the fifthembodiment, illustrated in FIG. 6, when a voltage of the variablevoltage source VV is varied, a DC current running into drains of then-type MOSFET transistors Q3 and Q4 varies, and further, a sourcevoltage of the n-type MOSFET transistors Q1 and Q2 also varies. Sincethe mutual conductance gin of the n-type MOSFET transistors Q1 and Q2varies in proportion with the voltage Vgs applied across a gate and asource, the mutual conductance gm₀ of the n-type MOSFET transistors Q1and Q2 in the equation (7) is not constant, but varies in accordancewith a voltage of the variable voltage source VV. If the mutualconductance gm₀ of the n-type MOSFET transistors Q1 and Q2 is notconstant, a voltage-current converting circuit (gm amplifier)unavoidably has a complex structure. In addition, each of the MOSFETtransistors may operate in an unsaturated area in accordance with avoltage of the variable voltage source VV.

In contrast, in the sixth embodiment, the p-type MOSFET transistors Q6and Q7 are electrically connected to the sources of the n-type MOSFETtransistors Q1 and Q2, and a bias voltage associated with a voltage ofthe variable voltage source VV, provided from the bias circuit 1, isapplied to the gates of the p-type MOSFET transistors Q6 and Q7. Thisensures that fluctuated DC current is compensated for. Thus, a DCvoltage at the sources of the n-type MOSFET transistors Q1 and Q2 isconstant independently of a voltage of the variable voltage source VV,and hence, the mutual conductance gm₀ of the n-type MOSFET transistorsQ1 and Q2 is also constant.

FIG. 9 is a circuit diagram of an example of the bias circuit in thesixth embodiment of the present invention, together with a circuitdiagram of an example of the bias circuit 1.

As illustrated in FIG. 9, the bias circuit 1 is comprised, for instance,of a p-type MOSFET transistor Q8, a n-type MOSFET transistor Q3 a, an-type MOSFET transistor Q1 a, a positive resistor R1 a, a variablevoltage source VVa, and a constant voltage source VS.

A gate and a drain in the p-type MOSFET transistor Q8 areshort-circuited with each other, and are electrically connected to anoutput terminal 1A of the bias circuit 1 and a source of the n-typeMOSFET transistor Q3 a. The n-type MOSFET transistor Q3 a has a drainelectrically connected to the variable voltage source VVa, a sourceelectrically connected to a gate and a source of the p-type MOSFETtransistor Q8, and a gate electrically connected to a junction node atwhich the n-type MOSFET transistor Q1 a and the positive resistor R1 aare electrically connected to each other. The variable voltage sourceVVa is electrically connected at one end thereof to a drain of then-type MOSFET transistor Q3 a, and at the other end grounded. The n-typeMOSFET transistor Q1 a has a gate electrically connected to the variablevoltage source VVa, and a source electrically connected to a gate of then-type MOSFET transistor Q3 a and the positive resistor R1 a. Thepositive resistor R1 a is electrically connected at one end thereof to agate of the n-type MOSFET transistor Q3 a and a source of the n-typeMOSFET transistor Q1 a, and at the other end grounded.

The n-type MOSFET transistor Q1 a, the n-type MOSFET transistor Q3 a,the positive resistor R1 a, and the variable voltage source VVa all inthe bias circuit 1 correspond to the n-type MOSFET transistor Q1, then-type MOSFET transistor Q3, the positive resistor R1, and the variablevoltage source VV all in the voltage-current converting circuit inaccordance with the sixth embodiment, illustrated in FIG. 8. A currentrunning across a drain and a source of the n-type MOSFET transistor Q3 ais identical with the same in the n-type MOSFET transistor Q3.

The constant voltage source VS providing a voltage (Vin+−Vin−)/2 iselectrically connected to a gate of the n-type MOSFET transistor Q1 a.

A source of the p-type MOSFET transistor Q8 having a gate and a drainshort-circuited with each other is electrically connected to a drain ofthe n-type MOSFET transistor Q3 a. A gate voltage of the p-type MOSFETtransistor Q8 is applied to gates of the n-type MOSFET transistors Q6and Q7 as a bias voltage.

In the voltage-current converting circuit illustrated in FIG. 9, when avoltage of the variable voltage source VV varies, a current runningthrough the n-type MOSFET transistors Q3 and Q4 varies, and a voltage ofthe variable voltage source VVa also varies. Hence, a currentfluctuation in the n-type MOSFET transistors Q3 and Q4 reflects on acurrent fluctuation in the n-type MOSFET transistor Q3 a, andaccordingly, on a current fluctuation in the p-type MOSFET transistorQ8.

Since the p-type MOSFET transistor Q8 and the p-type MOSFET transistorsQ6 and Q7 define a current mirror circuit, a current fluctuation in then-type MOSFET transistors Q3 and Q4 is applied to the n-type MOSFETtransistor Q3 a through the p-type MOSFET transistors Q6 and Q7.Accordingly, it is possible to keep a current running through n-typeMOSFET transistors Q1 and Q2 constant, even if a voltage of the variablevoltage source VV varies. Thus, it is possible to keep a source voltagein the n-type MOSFET transistors Q1 and Q2 constant, and hence, it isalso possible to keep the mutual conductance of the n-type MOSFETtransistors Q1 and Q2 constant.

Seventh Embodiment

FIG. 10 is a circuit diagram of a voltage-current converting circuit inaccordance with the seventh embodiment of the present invention.

The voltage-current converting circuit in accordance with the seventhembodiment additionally includes positive resistors R3 and R4 incomparison with the voltage-current converting circuit in accordancewith the fifth embodiment, illustrated in FIG. 6. Hence, parts orelements in FIG. 10 that correspond to those of FIG. 6 have beenprovided with the same reference numerals.

The positive resistor R3 is electrically connected in series between asource of the n-type MOSFET transistor Q1 and a junction node N1 atwhich the positive resistor R1, a drain of the n-type MOSFET transistorQ3, and a gate of the n-type MOSFET transistor Q4 are electricallyconnected to one another. The positive resistor R4 is electricallyconnected in series between a source of the n-type MOSFET transistor Q2and a junction node N2 at which the positive resistor R2, a drain of then-type MOSFET transistor Q4, and a gate of the n-type MOSFET transistorQ3 are electrically connected to one another.

In the voltage-current converting circuit in accordance with the fifthembodiment, illustrated in FIG. 6, the drains of the n-type MOSFETtransistors Q3 and Q4 each acting as a negative resistance device areelectrically connected to a junction node at which a source of then-type MOSFET transistor Q1 and the positive resistor R1 areelectrically connected to each other and a junction node at which asource of the n-type MOSFET transistor Q2 and the positive resistor R2are electrically connected to each other, respectively. In contrast, inthe seventh embodiment, the drains of the n-type MOSFET transistors Q3and Q4 each acting as a negative resistance device are electricallyconnected to the above-mentioned junction nodes N1 and N2, respectively.

The mutual conductance Gm of the voltage-current converting circuit inaccordance with the seventh embodiment is obtained by replacing R with(R_(R3)+1/(1/R_(R1)−gm_(Q3))) in the equation (1), wherein R_(R3)indicates a resistance of the positive resistor R3. That is, the mutualconductance Gm is equal to a sum of the mutual conductance of thevoltage-current converting circuit in accordance with the fifthembodiment, and the resistance R_(R3).

The voltage-current converting circuit in accordance with the seventhembodiment provides the same advantages as those obtained by thevoltage-current converting circuit in accordance with the firstembodiment. Furthermore, since the positive resistors R3 and R4 areelectrically connected between the sources of the n-type MOSFETtransistors Q1 and Q2, and the negative resistor NR, non-linearity ofthe n-type MOSFET transistors Q3 and Q4 is relaxed, ensuring that thevoltage-current converting circuit (gm amplifier) can accomplish morelinear operation.

Eighth Embodiment

FIG. 11 is a circuit diagram of a voltage-current converting circuit inaccordance with the eighth embodiment of the present invention.

The voltage-current converting circuit in accordance with the eighthembodiment includes p-type MOSFET transistors Q9 and Q10 in place of then-type MOSFET transistors Q3 and Q4 each acting as a negative resistancedevice, in comparison with the voltage-current converting circuit inaccordance with the fifth embodiment, illustrated in FIG. 6. Thevoltage-current converting circuit in accordance with the eighthembodiment is structurally identical with the voltage-current convertingcircuit in accordance with the fifth embodiment except theabove-mentioned difference. Hence, parts or elements in FIG. 11 thatcorrespond to those of FIG. 6 have been provided with the same referencenumerals.

The voltage-current converting circuit in accordance with the eighthembodiment in which electrical conductivity of the MOSFET transistorsdefining a negative resistance device is changed to p-type from n-typein comparison with the fifth embodiment provides the same advantages asthose obtained by the voltage-current converting circuit in accordancewith the fifth embodiment, illustrated in FIG. 6.

Ninth Embodiment

FIG. 12 is a circuit diagram of a voltage-current converting circuit inaccordance with the ninth embodiment of the present invention.

Whereas the voltage-current converting circuit in accordance with thefifth embodiment, illustrated in FIG. 6, is a differential circuit, thevoltage-current converting circuit in accordance with the ninthembodiment defines a single-end type gm amplifier. Hence, parts orelements in FIG. 12 that correspond to those of FIG. 6 have beenprovided with the same reference numerals. As an alternative, thevoltage-current converting circuit in accordance with the ninthembodiment has a negative resistance device NR having a differentstructure from that of the voltage-current converting circuit inaccordance with the first embodiment, illustrated in FIG. 1.

The voltage-current converting circuit in accordance with the ninthembodiment is comprised of an n-type MOSFET transistor Q1, a positiveresistor R1, and a resistor circuit.

The n-type MOSFET transistor Q1 receives an input voltage signal Vinthrough a gate thereof, and outputs an output current Iout. A source ofthe n-type MOSFET transistor Q1 is electrically connected to thepositive resistor R1 and the resistor circuit.

The positive resistor R1 is electrically connected at one end thereof toa source of the n-type MOSFET transistor Q1, and at the other endgrounded.

The resistor circuit is comprised of an n-type MOSFET transistor Q3acting as a negative resistance device, a phase-inverting circuit INV,and a variable voltage source VV.

The n-type MOSFET transistor Q3 has a drain electrically connected to ajunction node at which a source of the n-type MOSFET transistor Q1 andthe positive resistor R1 are electrically connected to each other, andfurther to an input terminal of the phase-inverting circuit INV, asource electrically connected to the variable voltage source VV, and agate electrically connected to an output terminal of the phase-invertingcircuit INV.

The phase-inverting circuit INV has an input terminal electricallyconnected to a junction node at which a drain of the n-type MOSFETtransistor Q3, a source of the n-type MOSFET transistor Q1, and thepositive resistor R1 are electrically connected to one another, and anoutput terminal electrically connected to a gate of the n-type MOSFETtransistor Q3.

The variable voltage source VV is electrically connected at one endthereof to a source of the n-type MOSFET transistor Q3, and at the otherend grounded.

A phase-inverted signal obtained by inverting a voltage signal of adrain of the n-type MOSFET transistor Q3 through the phase-invertingcircuit INV is input into a gate of the n-type MOSFET transistor Q3acting as a negative resistance device.

FIG. 13 is a circuit diagram of an example of the phase-invertingcircuit INV.

As illustrated in FIG. 13, the phase-inverting circuit INV is comprisedof p-type MOSFET transistors Q11, Q13 and n-type MOSFET transistors Q12,Q14.

The p-type MOSFET transistor Q11 and the n-type MOSFET transistor Q12define an inverter, and the p-type MOSFET transistor Q13 and the n-typeMOSFET transistor Q14 define an inverter-type load in which an inputterminal and an output terminal are short-circuited with each other.These two inverters are designed to have a theoretical threshold voltageequal to a DC bias at a junction node at which the positive resistor R1and a drain of the n-type MOSFET transistor Q3 are electricallyconnected to each other.

A negative resistance of the n-type MOSFET transistor Q3 is controlledby controlling a voltage of the variable voltage source VV to therebyvary a voltage between a source and a gate of the n-type MOSFETtransistor Q3.

Tenth Embodiment

FIG. 14 is a circuit diagram of a voltage-current converting circuit inaccordance with the tenth embodiment of the present invention.

The voltage-current converting circuit in accordance with the tenthembodiment does not include the variable voltage source VV, but includesvariable resistors R5 and R6 each having a positive resistance, in placeof the positive resistors R1 and R2, in comparison with thevoltage-current converting circuit in accordance with the fifthembodiment, illustrated in FIG. 6. The voltage-current convertingcircuit in accordance with the tenth embodiment is structurallyidentical with the voltage-current converting circuit in accordance withthe fifth embodiment except the above-mentioned difference. Hence, partsor elements in FIG. 14 that correspond to those of FIG. 6 have beenprovided with the same reference numerals.

In the voltage-current converting circuit in accordance with the fifthembodiment, illustrated in FIG. 6, a gain of the voltage-currentconverting circuit is controlled by controlling a negative resistance.In contrast, in the tenth embodiment, a gain of the voltage-currentconverting circuit is controlled by controlling the variable positiveresistors R5 and R6.

FIG. 15 is a circuit diagram of an example of the variable positiveresistors R5 and R6.

For instance, the variable positive resistor R5 or R6 is comprised of apositive resistor R7, and an n-type MOSFET transistor Q15 electricallyconnected in series to the positive resistor R7.

The n-type MOSFET transistor Q15 is used in an unsaturated area in whichVgs is greater than (Vds+Vth) so as to use the n-type MOSFET transistorQ15 as a resistor, wherein Vgs indicates a voltage between a gate and asource, Vds indicates a voltage between a drain and a source, and Vthindicates a threshold voltage of the n-type MOSFET transistor Q15. Aresistance of the n-type MOSFET transistor Q15 is controlled inaccordance with a bias voltage input into a gate of the n-type MOSFETtransistor Q15.

FIG. 16 is a circuit diagram of another example of the variable positiveresistors R5 and R6.

For instance, the variable positive resistor R5 or R6 is comprised of an-type MOSFET transistor Q16 in which a gate and a drain areshort-circuited with each other, and a variable voltage source VVelectrically connected at one end thereof in series to a source of then-type MOSFET transistor Q16, and at the other end grounded.

A positive resistance of the variable positive resistors R5 and R6 iscontrolled by controlling a voltage of the variable voltage source VV tothereby vary a voltage between a gate and a source in the n-type MOSFETtransistor Q16.

In the voltage-current converting circuit in accordance with the tenthembodiment, illustrated in FIG. 14, a voltage source providing aconstant voltage may be arranged between sources of the n-type MOSFETtransistors Q3 and Q4, and a grounded voltage.

Eleventh Embodiment

FIG. 17 is a circuit diagram of a voltage-current converting circuit inaccordance with the eleventh embodiment of the present invention.

The voltage-current converting circuit in accordance with the eleventhembodiment does not include the resistors R1 and R2, in comparison withthe voltage-current converting circuit in accordance with the fifthembodiment, illustrated in FIG. 6. The voltage-current convertingcircuit in accordance with the eleventh embodiment is structurallyidentical with the voltage-current converting circuit in accordance withthe fifth embodiment, illustrated in FIG. 6, except the above-mentioneddifference. Hence, parts or elements in FIG. 17 that correspond to thoseof FIG. 6 have been provided with the same reference numerals.

The mutual conductance Gm of the voltage-current converting circuit inaccordance with the eleventh embodiment is obtained by setting theresistance R_(R1) of the positive resistor R1 to be equal to infinite inthe equation (2).

The voltage-current converting circuit in accordance with the eleventhembodiment makes it possible to much vary the mutual conductance Gmthereof even by small fluctuation of a voltage provided from thevariable voltage source VV.

Twelfth Embodiment

FIG. 18 is a circuit diagram of a voltage-current converting circuit inaccordance with the twelfth embodiment of the present invention.

The voltage-current converting circuit in accordance with the twelfthembodiment does not include the resistors R1 and R2, in comparison withthe voltage-current converting circuit in accordance with the seventhembodiment, illustrated in FIG. 10. As an alternative, thevoltage-current converting circuit in accordance with the twelfthembodiment additionally includes positive resistors R3 and R4 incomparison with the voltage-current converting circuit in accordancewith the eleventh embodiment, illustrated in FIG. 17.

The positive resistor R3 is electrically connected in series between asource of the n-type MOSFET transistor Q1, and a drain of the n-typeMOSFET transistor Q3 and a gate of the n-type MOSFET transistor Q4. Thepositive resistor R4 is electrically connected in series between asource of the n-type MOSFET transistor Q2, and a drain of the n-typeMOSFET transistor Q4 and a gate of the n-type MOSFET transistor Q3.

The voltage-current converting circuit in accordance with the twelfthembodiment is structurally identical with either the voltage-currentconverting circuit in accordance with the seventh embodiment,illustrated in FIG. 10, or the voltage-current converting circuit inaccordance with the eleventh embodiment, illustrated in FIG. 17, exceptthe above-mentioned difference. Hence, parts or elements in FIG. 18 thatcorrespond to those of FIG. 10 or FIG. 17 have been provided with thesame reference numerals.

The mutual conductance Gm of the voltage-current converting circuit inaccordance with the twelfth embodiment is obtained by replacing R with(R_(R3)−1/gm_(Q3))) in the equation (1).

The voltage-current converting circuit in accordance with the twelfthembodiment provides the same advantages as those obtained by thevoltage-current converting circuit in accordance with the eleventhembodiment. Furthermore, since the positive resistors R3 and R4 areelectrically connected between the sources of the n-type MOSFETtransistors Q1 and Q2, and the negative resistor NR, non-linearity ofthe n-type MOSFET transistors Q3 and Q4 is relaxed, ensuring that thevoltage-current converting circuit can accomplish more linear operation.

Thirteenth Embodiment

FIG. 19 is a circuit diagram of a voltage-current converting circuit inaccordance with the thirteenth embodiment of the present invention.

The voltage-current converting circuit in accordance with the thirteenthembodiment includes npn-type bipolar transistors B1, B2, B3 and B4 inplace of the n-type MOSFET transistors Q1, Q2, Q3 and Q4, respectively,in comparison with the voltage-current converting circuit in accordancewith the fifth embodiment, illustrated in FIG. 6. The voltage-currentconverting circuit in accordance with the thirteenth embodiment isstructurally identical with the voltage-current converting circuit inaccordance with the fifth embodiment, illustrated in FIG. 6, except theabove-mentioned difference. Hence, parts or elements in FIG. 19 thatcorrespond to those of FIG. 6 have been provided with the same referencenumerals.

By defining “gm” as a voltage-current conversion gain of a bipolartransistor, the voltage-current converting circuit operates inaccordance with the equation (7), similarly to the voltage-currentconverting circuit in accordance with the fifth embodiment, illustratedin FIG. 6. It should be noted that “gm_(Q3)” in the equation (7) isreplaced with “gm_(B3)” defined as gm of the bipolar transistor B3.

The voltage-current converting circuit in accordance with the thirteenthembodiment provides the same advantages as those obtained by thevoltage-current converting circuit in accordance with the fifthembodiment. That is, a bipolar transistor may be used in place of aMOSFET transistor acting as an active device, in the above-mentionedfirst to twelfth embodiments.

Fourteenth Embodiment

FIG. 20 is a circuit diagram of a voltage-current converting circuit inaccordance with the fourteenth embodiment of the present invention.

The voltage-current converting circuit in accordance with the fourteenthembodiment includes a tunnel diode TD as a negative resistor NR, incomparison with the voltage-current converting circuit in accordancewith the first embodiment, illustrated in FIG. 1.

Specifically, the negative resistor in the voltage-current convertingcircuit in accordance with the fourteenth embodiment is comprised of atunnel diode TD having an input terminal electrically connected to ajunction node at which a source of the n-type MOSFET transistor Q1 andthe positive resistor R1 are electrically connected to each other, andan output terminal electrically connected to a later-mentioned variablevoltage source VV, and a variable voltage source VV electricallyconnected at one end thereof to the tunnel diode TD, and at the otherend grounded. The voltage-current converting circuit in accordance withthe fourteenth embodiment is structurally identical with thevoltage-current converting circuit in accordance with the firstembodiment, illustrated in FIG. 1, except the above-mentioned negativeresistor.

The variable voltage source VV is electrically connected between thetunnel diode TD and a ground. By controlling a bias voltage, a negativeresistance of the negative resistor NR can be controlled.

Fifteenth Embodiment

FIG. 21(a) is a circuit diagram of a filter circuit in accordance withthe fifteenth embodiment of the present invention.

The filter circuit illustrated in FIG. 21(a) is a secondary low-passfilter circuit having a variable bandwidth, comprising first to fourthvoltage-current converting circuits Gm₁, Gm₂, Gm₃ and Gm₄, and first andsecond capacity devices C₁ and C₂.

Each of two output terminals of the first voltage-current convertingcircuits Gm₁ is electrically connected to each of two input terminals ofthe second voltage-current converting circuits Gm₂, and each of twooutput terminals of the second voltage-current converting circuits Gm₂is electrically connected to both each of two input terminals of thethird voltage-current converting circuits Gm₃ and each of two inputterminals of the fourth voltage-current converting circuits Gm₄. Each oftwo output terminals of the third voltage-current converting circuitsGm₃ is electrically connected to each of two output terminals of thefourth voltage-current converting circuits Gm₄. That is, the thirdvoltage-current converting circuits Gm₃ and the fourth voltage-currentconverting circuits Gm₄ are electrically connected in parallel with eachother. Each of two input terminals of the second voltage-currentconverting circuits Gm₂ is electrically connected to each of two outputterminals thereof.

A variable voltage source VV is electrically connected to each of thefirst to fourth voltage-current converting circuits Gm₁, Gm₂, Gm₃ andGm₄.

The first capacity device C₁ is electrically connected between the twooutput terminals of the first voltage-current converting circuits Gm₁,and the second capacity device C₂ is electrically connected between thetwo output terminals of the fourth voltage-current converting circuitsGm₄.

FIG. 21(b) is a circuit diagram of the first to fourth voltage-currentconverting circuits Gm₁, Gm₂, Gm₃ and Gm₄.

As is obvious in view of FIG. 21(b), each of the first to fourthvoltage-current converting circuits Gm₁, Gm₂, Gm₃ and Gm₄ is comprisedof the voltage-current converting circuit in accordance with the fifthembodiment, illustrated in FIG. 6.

A transfer function of the filter circuit in accordance with thefifteenth embodiment is expressed with the equation (8). $\begin{matrix}{{F(s)} = \frac{\frac{{gm}_{1} \cdot {gm}_{3}}{C_{1} \cdot C_{2}}}{s^{2} + {\frac{{gm}_{2}}{C_{1}}s} + \frac{{gm}_{3} \cdot {gm}_{4}}{C_{1} \cdot C_{2}}}} & (8)\end{matrix}$

If total gains of the first to fourth voltage-current convertingcircuits Gm₁, Gm₂, Gm₃ and Gm₄ are multiplied by A by controlling avoltage provided from the variable voltage source VV, the transferfunction is expressed with the following equation. $\begin{matrix}{\frac{\frac{A \cdot {gm}_{1} \cdot A \cdot {gm}_{3}}{C_{1} \cdot C_{2}}}{s^{2} + {\frac{A \cdot {gm}_{2}}{C_{1}}s} + \frac{A \cdot {gm}_{3} \cdot A \cdot {gm}_{4}}{C_{1} \cdot C_{2}}} = \frac{\frac{{gm}_{1} \cdot {gm}_{3}}{C_{1} \cdot C_{2}}}{\left( \frac{s}{A} \right)^{2} + {\frac{{gm}_{2}}{C_{1}} \cdot \frac{s}{A}} + \frac{{gm}_{3} \cdot {gm}_{4}}{C_{1} \cdot C_{2}}}} \\{= {F\left( \frac{s}{A} \right)}}\end{matrix}$

The latter transfer function is scaled A times relative to the formertransfer function with respect to a frequency.

FIG. 22 is a graph showing a relation between a gain and a frequency inthe filter circuit in accordance with the fifteenth embodiment. Thesolid line 221 shows the relation obtained by the transfer functionexpressed with the equation (8), and the solid line 222 shows therelation obtained by the latter transfer function.

As illustrated in FIG. 22, with respect to a certain frequency F, abandwidth in the latter transfer function is amplified A times relativeto a bandwidth in the transfer function expressed with the equation (8).

Though each of the first to fourth voltage-current converting circuitsGm₁, Gm₂, Gm₃ and Gm₄ constituting the filter circuit in accordance withthe fifteenth embodiment is comprised of the voltage-current convertingcircuit in accordance with the fifth embodiment, illustrated in FIG. 6,the voltage-current converting circuits in accordance with the otherembodiments may be used.

Furthermore, it is not always necessary for the first to fourthvoltage-current converting circuits Gm₁, Gm₂, Gm₃ and Gm₄ to becomprised of the same voltage-current converting circuit. The first tofourth voltage-current converting circuits Gm₁, Gm₂, Gm₃ and Gm₄ may becomprised of different voltage-current converting circuits from oneanother. For instance, the first voltage-current converting circuits Gm₁may be comprised of the voltage-current converting circuit in accordancewith the fifth embodiment, the second voltage-current convertingcircuits Gm₂ may be comprised of the voltage-current converting circuitin accordance with the sixth embodiment, the third voltage-currentconverting circuits Gm₃ may be comprised of the voltage-currentconverting circuit in accordance with the seventh embodiment, and thefourth voltage-current converting circuits Gm₄ may be comprised of thevoltage-current converting circuit in accordance with the eighthembodiment.

While the present invention has been described in connection withcertain preferred embodiments, it is to be understood that the subjectmatter encompassed by way of the present invention is not to be limitedto those specific embodiments. On the contrary, it is intended for thesubject matter of the invention to include all alternatives,modifications and equivalents as can be included within the spirit andscope of the following claims.

For instance, one of positive and negative resistors is comprised of aresistor having a variable resistance, and the other is comprised of aresistor having a fixed resistance in the above-mentioned embodiments.Instead, both of positive and negative resistors may be comprised of aresistor having a variable resistance.

INDUSTRIAL APPLICABILITY

As having been explained so far, the voltage-current converting circuitin accordance with the present invention is comprised of an activedevice carrying out voltage-current conversion, and avariable-resistance circuit electrically connected in series to theactive device, and including a negative resistance device. Hence, thevoltage-current converting circuit makes it possible to vary a gainwithout necessity of using a switching circuit by applying a controlvoltage to a single control terminal (a control terminal of the activedevice).

Furthermore, the present invention makes it possible to vary a gain witha circuit having a simple structure and including a small number ofdevices, ensuring reduction in a chip size, and further ensuringproposal of a small-sized voltage-current converting circuit in a lowprice. The voltage-current converting circuit in accordance with thepresent invention accomplishes a multi-mode channel-selecting filteradapted to a plurality of communication processes, in a small chip-area,contributing to accomplishment of a multi-mode receiver having a smallchip-area.

1. A voltage-current converting circuit which outputs a current inaccordance with a voltage input thereto, comprising: an active devicehaving an input terminal, an output terminal, and a grounded terminal,and carrying out voltage-current conversion; and a resistor circuitelectrically connected in series to said active device through saidgrounded terminal of said active device, and controlling a conversiongain of said active device, said resistor circuit having a variableresistance, and including a negative resistance device.
 2. Thevoltage-current converting circuit as set forth in claim 1, wherein saidactive device is comprised of a pair of active devices each operatingdifferentially with each other, and each having an input terminal, anoutput terminal, and a grounded terminal, and carrying outvoltage-current conversion, said resistor circuit is comprised of a pairof resistor circuits each electrically connected in series to each ofsaid active devices through said grounded terminal of each of saidactive devices, and each controlling a conversion gain of each of saidactive devices, each of said resistor circuits having a variableresistance, and including a negative resistance device.
 3. Thevoltage-current converting circuit as set forth in claim 1, wherein saidnegative resistance device has a variable resistance.
 4. Thevoltage-current converting circuit as set forth in claim 1, wherein saidresistor circuit or each of said resistor circuits is comprised of: oneor a plurality of resistance device(s) electrically connected in seriesto said active device; and a negative resistance device electricallyconnected in parallel with at least one of said resistance device(s). 5.The voltage-current converting circuit as set forth in claim 1, whereinsaid resistor circuit or each of said resistor circuits is comprised ofa first circuit comprised of a resistance device and a negativeresistance device electrically connected in series to each other, saidfirst circuit being electrically connected in series to said activedevice.
 6. The voltage-current converting circuit as set forth claim 1,wherein said resistor circuit or each of said resistor circuits iscomprised of a first resistance device electrically connected in seriesto said active device, and a second circuit electrically connected inparallel with said first resistance device, said second circuit beingcomprised of a negative resistance device, and a second resistancedevice electrically connected in series to said negative resistancedevice.
 7. The voltage-current converting circuit as set forth in claim2, wherein said negative resistance device of said pair of resistancecircuits is comprised of a pair of active devices electrically connectedin cross to each other and operating differentially with each other, andeach receiving, as an input signal, a node signal either at a connectionnode at which said active device and said resistor circuit areelectrically connected to each other or at any connection node in saidresistor circuit.
 8. The voltage-current converting circuit as set forthin claim 1, wherein said negative resistance device is comprised of afield effect transistor or a bipolar transistor.
 9. The voltage-currentconverting circuit as set forth in claim 8, wherein a resistance of saidnegative resistance device is controlled by controlling either a sourcevoltage or an emitter voltage of said field effect transistor or bipolartransistor.
 10. The voltage-current converting circuit as set forth inclaim 9, further comprising a voltage-providing circuit electricallyconnected between a reference voltage point and either a source or anemitter of said field effect transistor or bipolar transistor, andwherein a resistance of said negative resistance device is controlled bycontrolling a voltage provided by said voltage-providing circuit. 11.The voltage-current converting circuit as set forth in claim 10, whereinsaid voltage-providing circuit is comprised of: an operational amplifierhaving a first input terminal, a second input terminal, and an outputterminal; and an active device, wherein a voltage-control signal isinput to said first input signal of said operational amplifier, an inputterminal of said active device is electrically connected to said outputterminal of said operational amplifier, and an output terminal of saidactive device is electrically connected to said second input terminal ofsaid operational amplifier.
 12. The voltage-current converting circuitas set forth in claim 9, wherein said negative resistance device iscomprised of a pair of field effect transistors or bipolar transistorsoperating differentially with each other, wherein sources or emitters ofsaid field effect transistors or bipolar transistors are electricallyconnected to each other.
 13. The voltage-current converting circuit asset forth in claim 1, further comprising a voltage-controllerelectrically connected to a connection node at which said active deviceand said resistor circuit are electrically connected to each other, forcontrolling a voltage of said connection node.
 14. The voltage-currentconverting circuit as set forth in claim 13, wherein saidvoltage-controller is comprised of an active device electricallyconnected between a reference voltage and said connection node, andhaving an input terminal to which a bias signal is input.
 15. Thevoltage-current converting circuit as set forth in claim 13, whereinsaid voltage-controller compensates for voltage fluctuation caused atsaid connection node by variance of a resistance of said negativeresistance device.
 16. The voltage-current converting circuit as setforth in claim 1, wherein said resistor circuit includes a variableresistor having a positive resistance.
 17. The voltage-currentconverting circuit as set forth in claim 16, wherein said variableresistor is comprised of an active device.
 18. The voltage-currentconverting circuit as set forth in claim 1, wherein said active deviceis comprised of a field effect transistor or a bipolar transistor. 19.The voltage-current converting circuit as set forth in claim 1, whereinsaid active device carrying out voltage-current conversion and saidactive device comprising said negative resistance device are comprisedof the same transistors having electrical conductivities different fromeach other.
 20. A filtering circuit including a combination circuitcomprised of the voltage-current converting circuit, and a capacitydevice, wherein a pass band is controlled by varying a gain of saidvoltage-current converting circuit, said voltage-current convertingcircuit comprising: an active device having an input terminal, an outputterminal, and a grounded terminal, and carrying out voltage-currentconversion; and a resistor circuit electrically connected in series tosaid active device through said grounded terminal of said active device,and controlling a conversion gain of said active device, said resistorcircuit having a variable resistance, and including a negativeresistance device.